Systems and methods for high precision cable length measurement in a communication system

ABSTRACT

Systems are provided for cable length measurement in a communication system. The systems include a transmitter, a receiver, a signal sampler and a cable length calculation unit. The transmitter is configured to transmit a plurality of data symbols at a first data rate via a wired data communication link, and the receiver is configured to receive a reflection signal. The signal sampler is configured to sample the received reflection signal using a phase shift number of shifting sampling phases to generate reflection samples, and combine the reflection samples with different sampling phases to generate a series of reflection samples corresponding to a second data rate higher than the first data rate. The cable length calculation unit is configured to determine a delay parameter from the series of reflection samples, and generate an estimate of a length of the data communication link.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a continuation of U.S. patent application Ser. No.15/470,655, filed Mar. 27, 2017 (now U.S. Pat. No. 10,190,862), which inturn claims the benefit under 35 U.S.C. § 119(e) of U.S. ProvisionalPatent Application No. 62/313,683, filed Mar. 25, 2016, which are herebyincorporated by reference herein in their entireties.

FIELD OF USE

This disclosure relates to performance measurement of a cablecommunication system, and specifically, a high precision cable lengthmeasurement mechanism.

BACKGROUND OF THE DISCLOSURE

The background description provided herein is for the purpose ofgenerally presenting the context of the disclosure. Work of theinventors hereof, to the extent the work is described in this backgroundsection, as well as aspects of the description that does not otherwisequalify as prior art at the time of filing, are neither expressly norimpliedly admitted to be prior art against the present disclosure.

High-speed Ethernet communication is commonly used for computernetworking and in recent years has been adapted for use in automotiveenvironments. For example, an automotive cable is used to connectsub-systems of an automobile, including for example engine, braking,steering, safety and various sensor sub-systems. Such cablecommunication connection is usually subject to stringent electromagneticinterference (EMI) requirements in order to provide safe and efficienttransfer of data among the various automobile sub-systems.Characteristics of the cable connection, such as the cable length, aremeasured to assess the performance of the connection.

Existing methods to measure a cable length include using time domainreflection (TDR) or digital signal processing (DSP) based echo response.In these methods, the measurement resolution is highly dependent on thesampling rate (usually equivalent to the data symbol rate) or the pulsewidth of the incident signal that is sent to generate a reflection orecho. When the sampling rate, i.e., the transmission data symbol rate,is low, the measurement resolution of the cable length is usuallyunsatisfactory.

SUMMARY

Embodiments described herein provide a method for cable lengthmeasurement in a communication system. At a transmitter, a plurality ofdata symbols are transmitted at a first data rate via a wired datacommunication link. In response to the transmitting, at a receiver, areflection signal is received from the wired data communication link. Aphase shift number of shifting sampling phases are obtained, at a signalsampler, for sampling the received reflection signal. The receivedreflection signal is sampled, at the sampler, based on the phase shiftnumber of shifting sampling phases to generate reflection samples. Eachreflection sample corresponds to a sampled value of the receivedreflection signal shifted by a sampling phase from the phase shiftnumber of shifting sampling phases. The reflection samples withdifferent sampling phases are combined to generate a series ofreflection samples corresponding to a second data rate higher than thefirst data rate. A delay parameter indicative of a transmission time fora signal to transmit from the transceiver to an end of the wired datacommunication link, is determined based on values of the series ofreflection samples. An estimate of a length of the data communicationlink is generated based at least in part on the determined delayparameter.

In some implementations, a first timestamp is detected corresponding toa center of a mass of the series of reflection samples. A secondtimestamp of a data symbol is identified corresponding to the center ofthe mass, and the delay parameter is generated by comparing the firsttimestamp and the second timestamp.

In some implementations, the estimate of the length of the datacommunication link is calculated by multiplying a constant signaltransmission speed with the delay parameter associated with theplurality of reflection response samples, wherein the estimate of thelength has a resolution that is substantially equivalent to a lengthestimate based on data transmission at the second data rate.

In some implementations, a phase of a data symbol is shifted from amongthe plurality of data symbols with the phase shift number of shiftingsampling phases to generate a series of data symbols. The series of datasymbols are transmitted at the second data rate.

In some implementations, reflection noise is removed from the pluralityof reflection samples. Reflection coefficients are adapted associatedwith the plurality of reflection samples. The plurality of reflectionsamples are combined to obtain a plurality of multi-phase reflectionresponse samples. The plurality of multi-phase reflection samples havethe second data rate.

In some implementations, a matching filter is applied to the reflectionsignal to reduce reflection tail noise. The matching filter increases asignal-to-noise ratio of the analog reflection signal.

In some implementations, a plurality of matching filter coefficients aredetermined based at least in part on a shape characteristic of thereflection signal at an end of the data communication link, a connectorreflection or a bus interface network of the wired data communicationlink.

In some implementations, the plurality of matching filter coefficientsare dynamically updated based at least in part on a shape of thereflection signal to reduce reflection tail noise.

In some implementations, the reflection signal is sampled based on a setof shift sampling phases different from the phase shift number of shiftsampling phases to obtain a different set of reflection samples. Adifferent estimate of the length of the wired data communication link isgenerated based at least in part on the different set of reflectionresponse samples.

In some implementations, the number of the set of shift sampling phasesis increased for sampling to increase measurement resolution of the setof shift sampling phases.

BRIEF DESCRIPTION OF THE DRAWINGS

Further features of the disclosure, its nature and various advantageswill become apparent upon consideration of the following detaileddescription, taken in conjunction with the accompanying drawings, inwhich like reference characters refer to like parts throughout, and inwhich:

FIG. 1 is a block diagram illustrating DSP echo response based cablemeasurement with phase shifting sampling, according to some embodimentsdescribed herein;

FIG. 2 is a block diagram illustrating TDR based cable measurement withphase shifting sampling and a matching filter (MF) to increasesignal-to-noise ratio (SNR) of the received reflection signal, accordingto some embodiments described herein;

FIGS. 3-4 provide data plot diagrams illustrating a reflection responsewith increased SNR after an MF is applied, according to some embodimentsdescribed herein; and

FIG. 5 provides an example logic flow diagram illustrating aspects ofmeasuring cable length with enhanced precision in a low data symbol rateenvironment, according to some embodiments described herein.

DETAILED DESCRIPTION

This disclosure describes methods and systems for providing a highprecision cable length measurement mechanism by shifting sampling phasesof a transmission signal to generate transmission symbols at a highersymbol rate.

In some embodiments, cable length measurement is conducted based onmeasuring the delay of a reflection signal in response to an incidentsignal that is sent from the transmitter. An incident signal, e.g.,either analog or digital, is usually sent from a transmitter onto acable connection. When the incident signal reaches the end of the cable,due to the impedance variation at the end of the cable, some of theincident signal will be reflected back to the transmitter in the form ofa reflection signal. The shape of the reflection signal is usuallysubstantially similar to the transmitted incident signal. The speed ofsignal propagation is almost constant for a given transmission medium,e.g., a copper or optical cable. Thus, the transmission delay of theincident signal to transmit from the transmitter to the end of thecable, and then reflect back from the end of the cable to thetransmitter, is usually measured to calculate an estimate of the cablelength. If the transmission delay is represented as τ, and the signaltravel speed υ is considered as a constant, an estimate of the cablelength d is calculated as d=υ τ/2. Thus, the measurement resolution ofthe delay parameter τ determines the measurement resolution of the cablelength d.

The transmission delay is usually measured by identifying signal peaksof the reflection signal. For example, a first timestamp of a signalpeak in the incident signal is recorded, and then the correspondingsignal peak in the reflection signal needs to be detected. A secondtimestamp associated with corresponding signal peak in the reflectionsignal is used to subtract the first timestamp to obtain thetransmission delay parameter τ.

In detecting a signal peak of the reflection signal, the sampling periodand/or the pulse width of the incident signal affect the measurementresolution, because a signal section of the entire pulse width cansometimes be identified as the “peak.” Thus when the data symbol rate isrelatively low, meaning the sampling period and the pulse width of theincident signal are relatively large, the “peak” section of the receivedreflection signal is in turn relatively wide, resulting in relativelylow precision in the detected “peak” of the received reflection signal.

For example, when a symbol rate of 66.7 mega-symbols per second (MSPS)is used in a 100BASE-T1 system, the measurement resolution of the delayparameter τ is 15 ns and thus the corresponding cable measurementresolution is around 1.5 m. In an automotive environment where the cablelength is usually less than 8 m, such measurement resolution of around1.5 m typically does not provide sufficient technical accuracy andprecision.

Embodiments described herein provide a phase shifting sampling mechanismto effectively increase the sampling rate or the data symbol rate, whichin turn enhances the measurement resolution of reflection delay, in asystem with a low symbol rate. A set of shifting sampling phases is usedto sample data symbols to be transmitted onto a cable connection, or areflection signal received from the cable connection. The set ofshifting sampling phases contains a number of shifting sampling phases,θ₁, θ₂, . . . , θ_(n), each representing a shifting phase of thesampling signal. Each sampling phase from the set of shifting samplingphases is used to sample the same signal. For example, to sample asampling signal s(2πt) with a shifting phase of θ₁, is similar to“shifting” the sampling signal along the time axis for an amount ofθ₁/2π, and taking a sample point at the time instant t. The resultingsignal sample is s[2π(t−θ₁)]. Thus, data symbols to be transmitted, or areceived signal, are to be sampled n times, each time with a samplingphase from the set of shifting sampling phases, θ₁, θ₂, . . . , θ_(n).By doing the multi-phase sampling, more data samples (with differentsampling phases) are generated as compared to only sampling the datasymbols or the received signal once.

In some implementations, data symbols to be transmitted are sampled andre-transmitted with the set of shifting sampling phases. For example,each data symbol for transmission is re-transmitted multiple times, eachtime with a shifting phase from the number of shifting sampling phases,θ₁, θ₂, . . . , θ_(n). Thus, as each data symbol is transmitted andrepeated in the form of a plurality of symbols with different shiftedphases, the data symbol rate is “virtually” increased, resulting in anarrower pulse width of the transmitted signal. Thus the reflectionsignal, in response to the transmitted signal, in turn has a narrowerpulse width for “peak” detection and thus a more precise measurement.

In another implementation, when a reflection or echo signal is receivedfrom the cable connection, in response to the transmission of the datasymbols, the reflection signal is sampled with the number of shiftingsampling phases, θ₁, θ₂, . . . , θ_(n), e.g., by time-shifting thereceived reflection signal according to each shifting sampling phase,and then taking a sample at a time instant. Thus, a series of reflectionsamples are generated, which has a quantity equivalent to reflectionsamples generated under a higher data symbol rate, e.g., similar to the“virtually” increased symbol rate as discussed above. With the“virtually” increased symbol rate and hence narrower pulse width,detection of the “peak” in the received reflection signal achievesenhanced resolution.

As a result, when either the transmitted symbols or the receivedreflection signal are sampled with the shifting sampling phases, themeasurement resolution of reflection delay is improved, and themeasurement resolution of the cable length is in turn improved as well.

FIG. 1 is a block diagram illustrating DSP echo response based cablemeasurement with phase shifting sampling, according to some embodimentsdescribed herein. The DSP echo response based cable measurementtechnique is implemented at a suitably configured DSP transceiver 100,in an embodiment. The DSP transceiver 100 is configured to obtain datasymbols 112 to transmit at a transmitter symbol generator 102, and inone implementation, all the data symbols 112 are converted to an analogsignal at a digital-to-analog converter 101. In another implementation,the data symbols are transmitted in a digital form. The data symbols orthe converted analog signal is then transmitted via a communicationlink, e.g., an Ethernet cable for example.

The DSP transceiver 100 also contains a receiver component where adigital phase controller 105 is configured to sample a received analogsignal, e.g., a reflection signal, at the analog-to-digital converter(ADC) 104. The shifting sampling phases generator 106 is configured togenerate a plurality of shifting sampling phases based on a phase shiftnumber n. In some implementations, the phase shift number n ispredefined. In some other implementations, the phase shift number n isdynamically adjusted to obtain different sets of shifting samplingphases. For example, when the phase shift number n=4, the shiftingsampling phases are 0, π/4, π/2, and 3π/4. Thus the resulting samples111 of the received signal, which is represented by r(t), are r[4k],r[4k+1], r[4k+2] and r[4k+3], k=0, 1, . . . , m−1, where m is themaximum number of reflection response taps along the cable, usuallydetermined by the cable length.

The output samples 111 with shifting phases from the ADC 104, r[4k],r[4k+1], r[4k+2] and r[4k+3], k=0, 1, . . . , m−1, are then processed tocancel echo noise by the echo canceller 103. For example, the echocanceller 103 is configured to, in one implementation, adopt common echosuppression or echo cancellation techniques. The transmission symbols112, after echo cancellation at the echo canceller 103, are subtractedfrom the output samples 111 at the logic operator 107, to generate asignal 113 with echo cancelled. The echo-cancelled signal 113 is thenprovided to an adaptation engine 108, at which the echo coefficients ofthe signal 113 are adapted accordingly. For example, in an embodiment,the adaptation engine 108 is configured to adapt the echo coefficientsto minimize the energy of the echo noise, e.g., by recursive leastsquare error (RLS) or the least mean squared error (LMS) methods. Afterecho adaption is performed at adaptation engine 108, the echo response109 is generated. For example, when the phase shift number n=4, and theshifting sampling phases are 0, n/4, n/2, and 3n/4, four groups of echoresponses 109 are generated, represented by ec(4k), ec(4k+1), ec(4k+2)and ec(4k+3), respectively, k=0, 1, . . . , m−1.

As the transmitter to receiver latency is fixed, in an embodiment, theecho responses with different phase shifts, e.g., ec(4k), ec(4k+1),ec(4k+2) and ec(4k+3), are combined to get a multi-phase reflection,which is equivalent to the echo responses with an n-times highersampling rate, where n is the phase shift number. Thus in the aboveexample, as each of the echo responses ec(4k), ec(4k+1), ec(4k+2) andec(4k+3) generates a delay measurement, the delay resolution is fourtimes more precise, e.g., the delay resolution after phase shiftingsampling=delay resolution without phase shifting sampling/4.

As such, when the delay measurement associated with the echo responses109 is used to calculate the cable length at the cable lengthcalculation unit 110, the cable length resolution is also four-timesmore precise in the above example as compared to measurement withoriginal received reflection signal without phase shifting sampling.

In another implementation, instead of sampling the received reflectionsignal with shifting sampling phases from a shifting sampling phasegenerator 106 at ADC 104, the transmission symbol generator 102 isconfigured to shift the phase of each data symbol. For example, for eachdata symbol t[n] and a set of shifting sampling phases 0, π/4, π/2, and3π/4, a sequence of data symbols is generated at the transmission symbolgenerator 102 via phase-shifting circuitry, i.e., t[4k], t[4k+1],t[4k+2], t[4k+3], . . . . Thus the set of resulting data symbols aftershifting phase sampling contains four times more data symbols totransmit within a same period of time, and thus can be considered as afour-times higher data symbol date. The data symbols are then passed onto the DAC 101 with different shifting phases to generate a transmissionsignal. In this way, the transmission signal with multi-phase sequenceyields a higher symbol rate. Specifically, with the higher symbol rate,in an embodiment, the transmission signal has a smaller pulse width ascompared to an original transmission signal if no shifting phasesampling was performed at the transmission symbol generator 102. Inresponse to the higher symbol rate of the transmission signal, anyreflection signal would have a similarly smaller pulse width. As aresult, as discussed above, the resolution of detecting a “peak” of thereflection signal is enhanced in this way, and thus yields measurementof the reflection delay with better resolution.

FIG. 2 is a block diagram illustrating TDR based cable measurement withphase shifting sampling and a matching filter (MF) to increasesignal-to-noise ratio (SNR) of the received reflection signal, accordingto some embodiments described herein. The phase shifting sampling asshown in FIG. 1 can be applied to the TDR mechanism 200 as well. Asshown in FIG. 2, the TDR transceiver 201 is configured to send anincident signal 211, which is reflected at the end of the cable 202, andthe reflection signal 212 is transmitted back to the transceiver 201. Ashifting sampling phase generator 106, e.g., similar to 106 discussed inconnection with FIG. 1, is configured to generate a plurality ofshifting sampling phases, which are transmitted to the phase samplingunit 205 (e.g., similar to 106 in FIG. 1) is configured to sample thereflection signal 212 with a plurality of shifting sampling phases. Forexample, if the reflection signal 212 is represented as r(t), withexample shifting sampling phases 0, π/4, π/2, and 3π/4, the reflectionsamples are r[4k], r[4k+1], r[4k+2] and r[4k+3], k=0, 1, . . . , m−1,where m is the maximum tap number. In this way, the generated reflectionresponse samples r[4k], r[4k+1], r[4k+2] and r[4k+3], k=0, 1, . . . ,m−1, are four-times “denser” than response samples generated withoutshifting sampling phases, e.g., equivalent to sampling the receivedreflection signal with only a phase of 0. The resulting reflectionresponse samples r[4k], r[4k+1], r[4k+2] and r[4k+3], k=0, 1, . . . ,m−1, thus have an equivalent data rate to reflection response samplesobtained at a four-times higher data symbol rate, as described inconnection with FIG. 1. Thus, with the four-times higher data symbolrate, a pulse width corresponding to the reflection samples withmulti-phases is reduced to be ¼ of the pulse width of the originalreflection signal. The measurement resolution of a “peak” of thereflection samples is thus enhanced by four times, and thus measurementresolution of the transmission delay and the cable length are enhancedby four times. Similarly, if the reflection signal 212 is sampled with nshifting phases, e.g., the reflection samples are r[nk], r[nk+1],r[nk+2] . . . r[nk+n−1], k=0, 1, . . . , m−1, where m is the maximum tapnumber, the measurement resolution of cable length is enhanced n times.

In some implementations, a MF 204 is disposed at the receiver, the MFbeing configured to process the reflection signal 212 before thereflection signal 212 is sampled at the phase sampling unit 205. Forexample, when the far end of the cable 202 is properly terminated, thereis little reflection and the remaining incident signal 211 is absorbedat the end of the cable 202 by termination. In this case, the receivedreflection signal 212 is too weak to detect, and the reflectiondetection can be unreliable due to the reduced signal strength andrelatively significant noise. The MF 204 is then used to suppress thenoise of the detection of the tail reflection and boost the SNR, in anembodiment.

In some implementations, the MF 204 is configured as a linear filterwith coefficients derived by the least squares residual method. Forexample, the MF coefficients are determined based on the reflectionresponse shape of the cable end, which includes the connector reflectionand the Bus Interface Network (BIN). For example, if the terminalreflection response is es(k), k=0, 1, . . . , p−1, where p is a shapingcharacteristic that indicates the length of the reflection signal in thetime-domain, e.g., a shaping the time-duration of the reflection signalreflected from the end of the cable 202, the MF coefficients aredetermined to be mf(k)=es(−k). Also, the MF is configured to representthe characteristics of the termination impedance at the end of the cable(e.g., 202 in FIG. 2), which is configured to separate the end of thecable and in-line connectors. In some implementations, the MF isconfigured to be a simple linear filter with the coefficients discussedabove to reduce hardware implementation complexity. In someimplementations, the MF coefficients are configured to be dynamicallyadjusted depending on characteristics of the received reflection signal,in order to achieve a desirable SNR.

FIGS. 3-4 provide data plot diagrams illustrating a reflection responsewith increased SNR after an MF is applied, according to some embodimentsdescribed herein. As shown in data plot diagram 300, the reflectionresponse signal (e.g., 212 in FIG. 2) after the MF (e.g., 204 in FIG. 2)is applied (as illustrated by the signal plot 301), has a greateramplitude than the original reflection response signal without the MF(as illustrated by the signal plot 302). Thus the peaks of thereflection response signal 301 are more identifiable by detecting thegreatest absolute value of the amplitude. An enlarged view 305 of a peakof the reflection signal is provided in FIG. 4. In some implementations,the center of mass of the reflection response signal is measured insteadof the peak of the reflection response signal, as the center of mass ofthe signal is less noisy to detect and with lower measurement error,especially with limited response tap number.

FIG. 5 provides an example logic flow diagram illustrating aspects ofmeasuring cable length with enhanced precision in a low data symbol rateenvironment, according to some embodiments described herein. At 501, anumber of data symbols are transmitted to a cable connection at a lowsymbol rate, e.g., at 66.7 MSPS in a 100BASE-T1 system. At 502, areflection signal (e.g., 212 in FIG. 2) is received from the cableconnection, in response to the transmission of the data symbols. At 503,a phase shift number, e.g., n, of shifting sampling phases are obtainedfor sampling. For example, when n=4, four shifting sampling phases of 0,π/4, π/2, and 3π/4 are to be used. At 504, the received reflectionsignal is sampled with the shifting sampling phases. For example, whenthe four shifting sampling phases of 0, π/4, π/2, and 3π/4 are used,four groups of reflection samples, e.g., ec[4k), ec[4k+1], ec[4k+2] andec[4k+3], respectively, k=0, 1, . . . , m−1, are obtained using each ofthe four shifting sampling phases. At 505, the reflection samplesobtained with different shifting sampling phases are combined to form aseries of reflection samples, e.g., ec(0), ec(1), ec(2), ec(3), ec(4),ec(5), ec(6), ec(7), . . . , ec(4m−4), ec(4m−3), ec(4m−2) and ec(4m−1),which is similar to the reflection samples obtained at a four-timeshigher symbol rate. Thus at 506, the delay parameter is calculated basedon the combined reflection samples, e.g., by detecting and comparing thetimestamp of a peak, or the center of the mass of the reflection samplesand the timestamp of a peak, or the center of the mass of thetransmitted data symbols. The detection of a peak, or the center of themass of the reflection samples is shown in FIGS. 3-4. At 507, anestimate of the cable length is calculated based on the delay parameterand a constant signal propagation speed of the Ethernet cableconnection.

In some implementations, a different set of shifting sampling phases isused to obtain a different set of reflection samples. For example, theshifting sampling phases are configured to change from 0, π/4, π/2, aπd3π/4, to 0, π/6, π/3, π/2, 2π/3 and 5π/6. A different series ofreflection samples is then generated based on the shifting samplingphases, and thus a different estimate of the length of the cable is inturn generated. The set of shifting sampling phases is configured to bedynamically changed to adapt to different measurement precisionrequirement, e.g., when the phase shifting number increases, themeasurement precision is usually enhanced accordingly.

While various embodiments of the present disclosure have been shown anddescribed herein, such embodiments are provided by way of example only.Numerous variations, changes, and substitutions relating to embodimentsdescribed herein are applicable without departing from the disclosure.It is noted that various alternatives to the embodiments of thedisclosure described herein may be employed in practicing thedisclosure. It is intended that the following claims define the scope ofthe disclosure and that methods and structures within the scope of theseclaims and their equivalents be covered thereby.

While operations are depicted in the drawings in a particular order,this is not to be construed as requiring that such operations beperformed in the particular order shown or in sequential order, or thatall illustrated operations be performed, to achieve the desirableresults.

The subject matter of this specification has been described in terms ofparticular aspects, but other aspects can be implemented and are withinthe scope of the following claims. For example, the actions recited inthe claims can be performed in a different order and still achievedesirable results. As one example, the process depicted in FIG. 10 doesnot necessarily require the particular order shown, or sequential order,to achieve desirable results. In certain implementations, multitaskingand parallel processing may be advantageous. Other variations are withinthe scope of the following claims.

What is claimed is:
 1. A method for cable length measurement in acommunication system, the method comprising: transmitting, at atransmitter, a plurality of data symbols at a first data rate via awired data communication link; in response to the transmitting,receiving, at a receiver, a reflection signal from the wired datacommunication link; sampling, at a signal sampler, the receivedreflection signal based on a phase shift number of shifting samplingphases; generating, from the sampling, a plurality of reflectionsamples, wherein the plurality of reflection samples corresponds to asecond data rate higher than the first data rate; determining, at acable length calculation unit, a delay parameter based on the pluralityof reflection samples corresponding to the second data rate; andgenerating, at the cable length calculation unit, an estimate of alength of the wired data communication link based at least in part onthe determined delay parameter.
 2. The method of claim 1, whereinsampling, at a signal sampler, the received reflection signal based on aphase shift number of shifting sampling phases comprises: for eachsampled value of the received reflection signal, generating acorresponding set of phase-shifted samples, wherein each of thephase-shifted samples is the sampled value shifted by a sampling phasefrom the phase shift number of shifting sampling phases; and combiningeach set of phase-shifted samples to generate the plurality ofreflection samples.
 3. The method of claim 1, wherein the delayparameter is indicative of a transmission time for a signal to transmitfrom the transmitter to an end of the wired data communication link. 4.The method of claim 1, further comprising: for each data symbol of thetransmitted plurality of data symbols, generating a phase shift numberof data symbols with different shifted phases corresponding to the phaseshift number of shifting sampling phases, and for each respective datasymbol of the phase shift number of data symbols, subtracting therespective data symbol from a respective reflection sample of the phaseshift number of reflection samples, respectively, to obtain a pluralityof reflection response samples at the second data rate; and whereindetermining, at the cable length calculation unit, the delay parameterbased on the plurality of reflection samples comprises determining thedelay parameter based on values of the plurality of reflection responsesamples.
 5. The method of claim 4, further comprising: detecting a firsttimestamp corresponding to a center of a mass of the plurality ofreflection response samples; identifying a second timestamp of a datasymbol corresponding to the center of the mass; and generating the delayparameter by comparing the first timestamp and the second timestamp. 6.The method of claim 1, further comprising: calculating the estimate ofthe length of the data communication link by multiplying a constantsignal transmission speed with the delay parameter, wherein the estimateof the length has a resolution that is substantially equivalent to alength estimate based on data transmission at the second data rate. 7.The method of claim 1, further comprising: shifting a phase of a datasymbol from among the plurality of data symbols with the phase shiftnumber of shifting sampling phases to generate a series of data symbols;and transmitting the series of data symbols at the second data rate. 8.The method of claim 1, further comprising: applying a matching filter tothe reflection signal to reduce reflection tail noise, wherein thematching filter increases a signal-to-noise ratio of the analogreflection signal.
 9. The method of claim 8, further comprising:determining a plurality of matching filter coefficients based at leastin part on a shape characteristic of the reflection signal at an end ofthe data communication link, a connector reflection, or a bus interfacenetwork of the wired data communication link.
 10. The method of claim 9,further comprising: dynamically updating the plurality of matchingfilter coefficients based at least in part on a shape of the reflectionsignal to reduce reflection tail noise.
 11. A system for cable lengthmeasurement in a communication system, the system comprising: atransmitter configured to transmit a plurality of data symbols at afirst data rate via a wired data communication link; a receiverconfigured to, in response to the transmitting, receive a reflectionsignal from the wired data communication link; a signal samplerconfigured to: sample the received reflection signal based on a phaseshift number of shifting sampling phases; generate, from the sampling, aplurality of reflection samples, wherein the plurality of reflectionsamples corresponds to a second data rate higher than the first datarate; and a cable length calculation unit configured to: determine adelay parameter based on the plurality of reflection samplescorresponding to the second data rate; and generating, at the cablelength calculation unit, an estimate of a length of the wired datacommunication link based at least in part on the determined delayparameter.
 12. The system of claim 11, wherein the signal sampler isfurther configured to sample the received reflection signal based on aphase shift number of shifting sampling phases by: for each sampledvalue of the received reflection signal, generate a corresponding set ofphase-shifted samples, wherein each of the phase-shifted samples is thesampled value shifted by a sampling phase from the phase shift number ofshifting sampling phases; and combine each set of phase-shifted samplesto generate the plurality of reflection samples.
 13. The system of claim11, wherein the delay parameter is indicative of a transmission time fora signal to transmit from the transmitter to an end of the wired datacommunication link.
 14. The system of claim 11, further comprising: areflection canceller configured to: for each data symbol of theplurality of data symbols, generating a phase shift number of datasymbols with different shifted phases corresponding to the phase shiftnumber of shifting sampling phases, and for each respective data symbolof the phase shift number of data symbols, subtracting the respectivedata symbol from a respective reflection sample from the phase shiftnumber of reflection samples, respectively, to obtain a plurality ofreflection response samples at the second data rate; and wherein thecable length calculation unit is further configured to determine thedelay parameter based on values of the plurality of reflection responsesamples.
 15. The system of claim 14, wherein the cable length calculatoris further configured to: detect a first timestamp corresponding to acenter of a mass of the plurality of reflection response samples;identify a second timestamp of a data symbol corresponding to the centerof the mass; and generate the delay parameter by comparing the firsttimestamp and the second timestamp.
 16. The system of claim 11, whereinthe cable length calculator is further configured to: calculate theestimate of the length of the data communication link by multiplying aconstant signal transmission speed with the delay parameter, wherein theestimate of the length has a resolution that is substantially equivalentto a length estimate based on data transmission at the second data rate.17. The system of claim 11, wherein the transmitter is furtherconfigured to: shift a phase of a data symbol from among the pluralityof data symbols with the phase shift number of shifting sampling phasesto generate a series of data symbols; and transmit the series of datasymbols at the second data rate.
 18. The system of claim 11, furthercomprising: a matching filter configured to apply to the reflectionsignal to reduce reflection tail noise, wherein the matching filterincreases a signal-to-noise ratio of the analog reflection signal. 19.The system of claim 18, wherein the matching filter includes a pluralityof matching filter coefficients determined based at least in part on ashape characteristic of the reflection signal at an end of the datacommunication link, a connector reflection, or a bus interface networkof the wired data communication link.
 20. The system of claim 19,wherein the plurality of matching filter coefficients are dynamicallyupdated based at least in part on a shape of the reflection signal toreduce reflection tail noise.